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RT9206
High Efficiency, Synchronous Buck with Dual Linear Controllers
General Description
The RT9206 is a low cost, combo power controller, which integrates a synchronous step-down voltage-mode PWM and two HV linear controllers. Directly drive external N-MOSFET makes it easy to implement a high efficiency and cost attractive power solution. Voltage mode control loop and constant operation frequency with external compensation network provide better stability in wide operation range. Adjustable operation frequency up to 600kHz can minimize the inductor size and PCB space. It is particularly suitable in wide input voltage range (from 4.75V to 28V) and multi-output applications. Linear controller features flexible linear power design. Delivered power can be simply decided by external N-MOSFET selection. Output voltage level is chosen via external resistor divider. The 0.8V internal reference can satisfy most of the applications. Under voltage lockout provide cost effective protection of output. RT9206 provides complete safety protection function: soft start, over current protection, over voltage and under voltage protection. Set current limit by choosing different MOSFET. Synchronous Buck control mode provides excellent over voltage protection by turning on low side MOSFET to prevent any damage of end device from abnormal voltage stress as over voltage condition occurs.
Features
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Wide Input Range (4.75V to 28V) 0.8V Internal Reference High Efficiency Synchronous Buck Topology Integrate two HV Linear Controllers Low cost N-MOSFET Design Duty Cycle from 0% to 90%. Adjustable switching frequency from 200kHz to 600kHz, Default 200kHz Sense OCP by low Side MOSFET RDS(ON) Power Good Signal Output RoHS Compliant and 100% Lead (Pb)-Free
Applications
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LCD Monitor Desk Note IEEE1394 Client Desktop IA Broadband
Pin Configurations
(TOP VIEW)
LDRV1 VDD LDRV2 LFB2 COMP FB PGOOD SS/EN 2 3 4 5 6 7 8 16 15 14 13 12 11 10 9 LFB1 BOOT UGATE PHS VINT LGATE GND RT
Ordering Information
RT9206 Package Type S : SOP-16 Operating Temperature Range P : Pb Free with Commercial Standard G : Green (Halogen Free with Commercial Standard) Note : RichTek Pb-free and Green products are : }RoHS compliant and compatible with the current requirements of IPC/JEDEC J-STD-020. }Suitable for use in SnPb or Pb-free soldering processes. }100%matte tin (Sn) plating.
SOP-16
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RT9206
Typical Application Circuit
R1 10 C1 1uF C15 500pF R2 11k VOUT1 VOUT2 3.3V Q3 C16 500pF C6 10uF VOUT3 2.5V C7 10uF Q4 R3 11k R11 5.6k C14 33nF R12 4.3K 390pF C13 R4 3.3k D11N4148 15 RBOOT 0 R13 C2 1uF C3 220uF VIN 12V
1 2 3 4 5 6 7 8 R10 51k
16 LDRV1 LFB1 RT9206CS 15 VDD BOOT 14 LDRV2 UGATE 13 LFB2 PHS 12 COMP VINT 11 FB LGATE 10 PGOOD GND 9 SS/EN RT Q5 EN
C9 1uF Si4800BDY C10 Q1 L1 1000uF 4.7uH Q2 C17 1000uF C11 1uF VOUT1 5V
C8 4.7uF 0 R14
RRT R8 3k R9 560
1uF CSS
PGOOD
Figure 1. Typical Application for 12V Input
R1 10 C1 1uF C15 500pF R2 11k Q3 C16 500pF C6 10uF V OUT3 2.5V C7 10uF Q4 R3 11k R11 5.6k C14 33nF R12 8.2K 220pF C13 R4 3.3k D11N4148 15 RBOOT 0 R13 C2 1uF C3 220uF
VIN 24V
VOUT1 VOUT2 3.3V
1 2 3 4 5 6 7 8 R10 51k
16 LDRV1 LFB1 RT9206CS 15 VDD BOOT 14 LDRV2 UGATE 13 LFB2 PHS 12 VINT COMP 11 FB LGATE 10 PGOOD GND 9 RT SS/EN Q5 EN
C9 1uF Si4800BDY Q1 L1 C10 1000uF 33uH Q2 C11 1uF VOUT1 5V
C8 4.7uF 0 R14
RRT R8 3k R9 560
4.7uF CSS
PGOOD
Figure 2. Typical Application for 24V Input
Note : R BOOT is a must to suppress ringing spike. www.richtek.com 2 DS9206-11 March 2007
RT9206
Function Block Diagram
LFB1 0.8V LFB2 VDD 0.6V + UV 0.6V 0.6V COMP 0.8V FB + GM RAMP Generator + OVP PG + OSC RT + PWM - CP
Soft Start
0.8V LCTR2 + +
+ +
LCTR1
LDRV1 LDRV2
6.0V Reg Thermal Protection + UV + UV Driver Control Logic
VINT
BOOT UGATE PHS VINT LGATE GND
SS/EN
8uA
PGOOD
1V
0.72V
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RT9206
Operation
Introduction The RT9206 is a combo controller, which integrates an adjustable frequency, voltage mode synchronous step down controller and two HV linear controllers. The synchronous step down controller consists of an internal precision reference, an internal oscillator, an error amplifier, a PWM comparator, control logic and floating gate driver, a programmable soft-start, a power good indicator, an over voltage protection, an over temperature protection and short circuit protection. The output voltage of the synchronous converter is set and controlled by the output of the error amplifier, which is the amplified error signal from the sensed output voltage and the voltage on non-inverting input, which is connected with internal 0.8V reference voltage. The amplified error signal is compared to a fixed frequency linear sawtooth ramp and generates fixed frequency pulse of variable dutycycle, which drivers the two N-Channel external MOSFETs. The timing of the synchronous converter is provide through an internal oscillator circuit and can be programmed between 200kHz to 600kHz via an external resistor connected between RT pin and ground. Soft-Start RT9206 has a programmable soft-start to control the output voltage rise time and limit the current surge at the startup. The soft-start will begin while VDD rises above POR threshold for correct start-up. Soft-start function operates by an internal sourcing current to charge an external capacitor to around the voltage of VINT. The soft-start signal, SS pin, is the third input non-inverting input of the PWM comparator. Before soft-start signal reach the bottom of the sawtooth ramp, inverting input of the PWM comparator, the soft-start current is twice of the normal soft-start current. Once the soft-start signal reach the bottom of the ramp, the soft-start current became normal, and start to increase duty cycle from zero to the point the feedback loop takes control. Power On Reset (POR) The power on reset circuit assures that the MOSFET driver outputs remain in the off state whenever the VDD supply voltages lower than the POR threshold. Over-Current Protection Whenever the over-current is occurred in soft-start or in normal operation period, It will shut down PWM signal, the MOSFET driver outputs remain in the off state, and latch soft-start signal low until restart VDD supply voltage. Over-Voltage Protection Once over-voltage protection occurred, it will turn on low side MOSFET and latch soft-start signal low to prevent end device form abnormal voltage stress. Restart VDD supply voltage will release the protection. Power Good Indicator The power good indicator is an open drain output to show whether the synchronous converter output ready or not. The power good indicator is available after soft-start end. Short-Circuit Protection The short-circuit phenomenon is sensed by the drop of output voltage, synchronous converter and two linear controller. Once the short-circuit occurred, the drop of output voltage lower than the under voltage threshold, 0.6V on feedback, the PWM signal will shut down and both of the external MOSFET will turn off and soft-start signal latch low. Soft-start signal, SS, is also connected to two linear controller error amplifier non-inverting input. Therefore, whenever the drop of output of the synchronous converter or two linear controllers lower than under voltage threshold, all MOSFET drivers will turn off.
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Pin Description
LDRV1(Pin 1) Linear controller 1 (LCTR1) driver. Connect to the gate of external N-Channel MOSFET pass transistor to form a positive linear regulator VDD (Pin 2) Input supply voltage LDRV2 (Pin 3) Linear controller 2 (LCTR2) driver. Connect to the gate of external N-Channel MOSFET pass transistor to form a positive linear regulator LFB2 (Pin 4) LDO2 feedback input. The feedback set point is 0.8V. Connect to a resistive divider between the positive linear regulator output and GND to adjust the output voltage. COMP (Pin 5) Switching regulator compensation pin. FB (Pin 6) Switching regulator feedback input. The feedback set point is 0.8V. Connect to a resistive divider between the switching regulator output and GND to adjust the output voltage. PGOOD (Pin 7) Open drain power good indicator. PGOOD is low when switching regulator output voltage is lower than 10% of its regulation voltage. Connect a pull high resistor between PGOOD and switching regulator output for pull high logic level voltage. SS/EN (Pin 8) Soft start input with 8uA sourcing current and IC enable control. RT (Pin 9) Operational frequency setting. Connect a resistor between RT and GND to set operational frequency. The operational frequency will nominally run at 200kHz when open. BOOT (Pin 15) High side floating driver supply with (+) terminal bootstrap flying capacitor connection. Voltage swing is from a diode drop below VINT to VIN+VINT LFB1 (Pin 16) LDO1 feedback input. The feed back set point is 0.8V. Connect to a resistive divider between the positive linear regulator output and GND to adjust the output voltage. PHS (Pin 13) Inductor connection with (-) terminal bootstrap flying capacitor connection. UGATE (Pin 14) High side gate driver. Drives high side N-Channel MOSFET with a voltage swing between BOOT and PHS LGATE (Pin 11) Low side gate driver. Drives low side N-MOSFET with a voltage swing between VINT and GND VINT (Pin 12) Internal 6.0V regulator output. The low side gate driver and control circuit and external bootstrap diode are powered by this voltage. Decouple this pin to power ground with a 4.7uF or greater ceramic capacitor close to the VINT pin. GND (Pin 10) Ground The formula between resistor setting and operational frequency are as follows: RRT = 62 x 10
8 3
FOSC - 200 x 10
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RT9206
Absolute Maximum Ratings
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(Note 1)
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Supply Voltage (VIN) ------------------------------------------------------------------------------------------------ -0.3 to 30V PHS --------------------------------------------------------------------------------------------------------------------- - 0.6V to 30V PHS (PHS Transient Time Interval < 50ns) --------------------------------------------------------------------- - 5V BOOT, UG to PHS --------------------------------------------------------------------------------------------------- - 0.3V to 7V BOOT to GND -------------------------------------------------------------------------------------------------------- - 0.3V to 35V LDRI1, LDRI2 --------------------------------------------------------------------------------------------------------- - 0.3V to 30V Power Good Voltage ------------------------------------------------------------------------------------------------- - 0.3V to 7V The other pins -------------------------------------------------------------------------------------------------------- - 0.3V to 7V Power Dissipation, PD @ TA = 25C SOP-16 ---------------------------------------------------------------------------------------------------------------- 0.625W Package Thermal Resistance SOP-16, JA --------------------------------------------------------------------------------------- 90C/W Junction Temperature ----------------------------------------------------------------------------------------------- 150C Lead Temperature (Soldering, 10 sec.) -------------------------------------------------------------------------- 260C Operation Temperature Range ------------------------------------------------------------------------------------- - 20C to 85C Storage Temperature Range --------------------------------------------------------------------------------------- - 65C to 150C ESD Susceptibility (Note 2) HBM (Human Body Mode) ----------------------------------------------------------------------------------------- 2kV MM (Machine Mode) ------------------------------------------------------------------------------------------------ 200V (Note 3)
Recommended Operating Conditions
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Ambient Temperature Range -------------------------------------------------------------------------------------- 0C to 70C Junction Temperature Range -------------------------------------------------------------------------------------- 0C to 125C
Electrical Characteristics
(VIN = 12V, FADJ left floating, TA = 25 C, Unless Otherwise specification)
Parameter System Supply Input Operation voltage Range Power On Reset Power On Reset Hysteresis Supply Current Shut Down Current Power Good under Threshold PG Fault Condition Soft Start Soft start Current Normal Operation Voltage Shut down Voltage
Symbol
Test Condition
Min
Typ
Max
Units
VDD POR
(Note 4)
4.75 3.8 200
--
28 4.7
V V mV mA mA % V
-1.3 1 ---
600 4 3.5 92 0.2
IDD IDD VFB VPG
VDD=30V, VSS=VINT VDD=30V, VSS<0.4V IPG= -4mA, VFB = 80%
--82 --
ISS VSS VSS
4 -0.4
8 VINT 0.7
12 ---
A V V
To be continued
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RT9206
Parameter PWM Section Reference Voltage Feedback Voltage Internal Voltage Internal Voltage Source Current PWM Section Oscillator Free Run Frequency Operation Frequency Setting Ramp Amplitude Maximum Duty Cycle Error Amplifier GM Compensation Source Current Compensation Sink Current Gate Driver Upper Gate Source (UGATE1 & 2) Upper Gate Sink (UGATE1 & 2) Lower Gate Source (LGATE1 & 2) Lower Gate Sink (LGATE1 & 2) Upper Gate Rising Time Upper Gate Falling Time Lower Gate Rising Time Lower Gate Falling Time Minimum On Time Protection Over Current Threshold Over Voltage Protection Under Voltage Protection Linear Controller Section Error Amplifier Feedback Voltage Output Current Protection Under Voltage Protection Over Temperature Protection LFB1 / LFB2 0.54 125 0.6 170 0.66 -V C LFB1 / LFB2 LDRV1/LDRV2 0.780 10 0.8 -0.824 -V mA VFB VFB -270 0.9 0.54 -300 1 0.6 -330 1.1 0.66 mV V V RUGATE RUGATE RLGATE RLGATE TR_UGATE TF_UGATE TR_LGATE TF_LGATE VDD = 12V, CLOAD = 3nF VDD = 12V, CLOAD = 3nF VDD = 12V, CLOAD = 3nF VDD = 12V, CLOAD = 3nF ---------5 5 3 1.5 30 30 30 30 -8 8 5 3 ----400 ns ns ns ns ns -45 45 1.6 90 90 -140 140 ms A A FOSC FOSC By setting RT (Note 5) 160 -30 -85 200 -1.9 90 240 +30 --kHz % V % VFB VINT IINT IINT = 10mA VIN = 12V 0.784 5.0 20 0.8 6 -0.816 6.5 -V V mA Symbol Test Condition Min Typ Max Units
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RT9206
Note 1. Stresses beyond those listed under "Absolute Maximum Ratings" may cause permanent damage to the device. These are stress ratings only, and functional operation of the device at these or any other conditions beyond those indicated in the operational sections of the specifications is not implied. Exposure to absolute maximum rating conditions for extended periods may affect device reliability. Note 2. Devices are ESD sensitive. Handling precaution recommended. The human body model is a 100pF capacitor discharged through a 1.5k resistor into each pin. Note 3. The device is not guaranteed to function outside its operating conditions. Note 4. V DD-V OUT2 or VDD-V OUT3 must be higher than 4V to keep linear controller operation Note 5. RRT =
62 x 10
8 3
Note 6. The LDOs are not suitable for low noise applications
FOSC - 200 x 10
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RT9206
Typical Operating Characteristics
Power On
IOUT1 = 4A IOUT2 = 0.5A IOUT3 = 0.5A 2V/Div 5V/Div 10V/Div
Power On
VIN
2V/Div
VIN VOUT1
2V/Div 2V/Div 2V/Div 5V/Div IOUT1 = 4A IOUT2 = 0.5A IOUT3 = 0.5A
VOUT1 VOUT3 VOUT2 PGOOD
VIN = 12V, f = 200kHz 2V/Div 5V/Div
VOUT2 VOUT3 PGOOD
VIN = 24V, f = 200kHz
Time (100ms/Div)
Time (200ms/Div)
Power Off
5V/Div 5V/Div IOUT1 = 4A IOUT2 = 0.5A IOUT3 = 0.5A 20V/Div
Power Off
IOUT1 = 4A IOUT2 = 0.5A IOUT3 = 0.5A
VIN
5V/Div
VIN VOUT1 VOUT2 VOUT3
5V/Div 2V/Div
VOUT1
2V/Div 2V/Div
VOUT3 VOUT2 PGOOD
2V/Div 5V/Div VIN = 24V, f = 200kHz
PGOOD
VIN = 12V, f = 200kHz
Time (20ms/Div)
Time (20ms/Div)
Bootstrap Wave Form
10V/Div 20V/Div
Bootstrap Wave Form
BOOT PHS
BOOT
10V/Div 5V/Div
PHS
10V/Div 20V/Div
UGATE UGATE
5V/Div
20V/Div
LGATE
VIN = 12V
LGATE
VIN = 24V
Time (1s/Div)
Time (1s/Div)
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RT9206
Dead Time
LGATE LGATE 2V/Div 2V/Div
Dead Time
UGATE 5V/Div
UGATE 10V/Div
VIN = 12V, IOUT1 = 4A
VIN = 24V, IOUT1 = 4A
Time (20ns/Div)
Time (20ns/Div)
Dead Time
UGATE 5V/Div UGATE 10V/Div
Dead Time
LGATE 2V/Div
LGATE 2V/Div
VIN = 12V, IOUT1 = 4A
VIN = 24V, IOUT1 = 4A
Time (20ns/Div)
Time (20ns/Div)
Dynamic Loading
Dynamic Loading
UGATE
UGATE
10V/Div
LGATE
10V/Div
LGATE
5V/Div VOUT1 100mV/Div IOUT1 5A/Div Time (10s/Div)
5V/Div
VOUT1 100mV/Div IOUT1 5A/Div
VIN = 12V
VIN = 12V
Time (10s/Div)
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RT9206
Short Latch
6.1
Soft Start vs. Temperature
VIN = 12V VIN = 12V f = 200kHz
UGATE
6.05
20V/Div
6
VSS (V)
5.95 5.9
2V/Div
VOUT1
10A/Div
5.85 5.8
IL Time (20s/Div)
-50 -25 0 25 50 75 100 125 150
Temperature (C)
Oscillator Frequency vs. Temperature
240 235 230 0.82
Reference Voltage vs. Temperature
0.815 0.81
VIN = 12V RT = floating
VIN = 12V f = 200kHz
Frequency (kHz)1
225 220 215 210 205 200 195 190 -50 -25 0 25 50 75 100 125 150
V REF (V)
0.805 0.8 0.795 0.79 0.785 -50 -25 0 25 50 75 100 125 150
Temperature (C)
Temperature (C)
POR(Rising/Falling) vs. Temperature
5 4.75 4.5 4.25 4 1055
Quiescent Current vs. Input Voltage
VSS = 0V
VIN = 12V f = 200kHz Rising
Quiescent Current (uA)
1050 1045 1040 1035 1030 1025
POR (V)
3.75 3.5 3.25 3 2.75 2.5 2.25 2 -50 -25 0 25 50 75 100 125 150
Falling
0
4
8
12
16
20
24
28
32
Temperature (C)
V IN (V)
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RT9206
Fosc vs. RRT
800
Efficiency vs. Load Current
96
VIN = 12V
A
700
94
Operation Frequency (kHz)
Efficiency (%)
600 500 400 300 200 100 0 0 5 10 15 20 25 30 35 40 45 50 55 60 65 70
92 90 88 86 84 82 80 0 1 2 3 4 5
VIN = 12V
VIN = 24V
VOUT = 5V Frequency = 200kHz
6 7
RRT (k)
Load Current (A)
Over Current Threshold vs. Temperature
450
VIN = 12V
Over Current Threshold (mV)1 (-mV)
400
350
300
250
200 -50 -25 0 25 50 75 100 125 150
Temperature (C)
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RT9206
Application Information
Synchronous Buck Converter The RT9206 is specifically designed for synchronous buck converter with wide input voltage from 4.75V to 28V and operating frequency from 200kHz to 600kHz. To fully utilize its advantages, peripheral components should be appropriately selected. The following information provides basic considerations for component selection. Output Inductor Selection The selection of output inductor is based on the considerations of efficiency, output power and operating frequency. Low inductance value has smaller size, but results in low efficiency, large ripple current and high output ripple voltage. Generally, an inductor that limits the ripple current (IL) between 20% and 50% of output current is appropriate. Figure 1 shows the typical topology of synchronous step-down converter and its related waveforms.
iS1 L + VL S1 S2 iS2 + V OR + VOC rC
+
is1
is2
Figure 1.The waveforms of synchronous step-down converter According to Figure 1 the ripple current of inductor can be calculated as follows :
VIN - VOUT = L IL t ; t = D fs ;D = VOUT VIN
L = (VIN - VOUT) x
VOUT VIN x fs x IL
(1)
iL
Where :
iC IOUT + RL
VIN= Maximum input voltage VOUT= Output Voltage
VIN
VOUT
t=S1 turn on time IL=Inductor current ripple f S= Switching frequency D=Duty Cycle
COUT -
Ts
rC= Equivalent series resistor of output capacitor
Vg1 Vg2 V IN -V OUT VL -V OUT Ton Toff
Output Capacitor Selection The selection of output capacitor depends on the output ripple voltage requirement. Practically, the output ripple voltage is a function of both capacitance value and the equivalent series resistance (ESR) rC. Figure 2 shows the related waveforms of output capacitor.
iL
IL=IOUT IL
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RT9206
iL
d iL VIN-VOUT =
dt L
d iL dt =
VOUT L
Input Capacitor Selection The selection of input capacitor is mainly based on its maximum ripple current capability. The buck converter draws pulsewise current from the input capacitor during the on time of S1 as shown in Figure 1. The RMS value of ripple current flowing through the input capacitor is described as :
Irms = IO D(1- D)
IO
TS
ic
1/2IL 0
IL
(A)
(6)
VOC
The input capacitor must be cable of handling this ripple current. Sometime, for higher efficiency the low ESR capacitor is necessarily.
VOC
Power MOSFET Selection
VOR
IL x rc
0
t1
t2
Figure 2. The related waveforms of output capacitor. The AC impedance of output capacitor at operating frequency is quite smaller than the load impedance, so the ripple current ( IL) of the inductor current flows mainly through output capacitor. The output ripple voltage is described as :
VOUT = VOR + VOC VOUT = IL x rC + 1 Co
The selection of MOSFETs is based on consideration of maximum gate-source voltage (Vgs), drain-source voltage (Vdss), maximum drain current (Id), drain-source on-state resistance Rds(on) and thermal management. The MOSFETs are driven by VINT that is internally regulated as 6.0V. Low threshold voltage MOSFET should be selected to guarantee that it could fully turn on at Vgs = 6.0V. The total power dissipation of external MOSFETs consists of conduction and switching losses. The conduction losses of high-side and low-side MOSFETs are described by equation (7) and (8), respectively. (High-side MOSFET)
2 PH - con = I0 x D x Rds(on) x r
(W)
(7)
(2)
(Low-side MOSFET)
2 PL - con = I0 x (1- D) x Rds(on) x r
(W)
(8)
ic d t
t2 t1
(3)
Where
V OUT = IL x rC +
1 VOUT 8 COL
r is temperature dependency of Rds(on)
(1- D)T
2 S
(4)
where VOR is caused by ESR and VOC by capacitance. For electrolytic capacitor application, typically 90~95% of the output voltage ripple is contributed by the ESR of output capacitor. So Equation (4) could be simplified as :
VOUT = IL x rC
The total switching loss is approximated as. Vds(off) Psw = IOUT x x (tr + tf) x fs (W) 2 Where
(9)
Vds(off) is voltage from drain to source at MOSFET off time. tr and tf are rise-time and fall-time, respectively. IOUT = Load current f s= Switching frequency
(5)
Users could connect capacitors in parallel to get calculated ESR.
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RT9206
The MOSFET should be capable of handling the power loss over the entire operating range. Design Example: Design the power stage for a synchronous step-down converter having the following specifications: VIN = 12V, VOUT = 5V, IOUT = 5A, VOUT < 25mV, switching frequency = 200kHz, to determine the value of inductor and output capacitor (Using electrolytic capacitor). First, select ripple current of inductor is 20% of output current, from equation (1)
L = (12 - 5) x 5 12 x 200K x 0.2 x 5 = 14.58 H
VOUT
Ra
(LFB1,LFB2)
FB
+ Rb 0.8V
Figure 3. The connected diagram of external voltage divider and reference voltage
Select L = 15H From equation (5) 25mV=1 x rC Select two electrolytic capacitors C = 470F,rC = 43m in parallel. Setting the Current Limit The RT9206 limits output current by sensing low side MOSFET voltage drop (VSD) when it turns on. The drop voltage caused by on-state resistance RDS(ON) is described as : VSD=RDS(ON) x IL (10)
If high value resistors are used, the input bias current of FB pin could cause a slight increase in output voltage. The output voltage set point can be more accurate by using precision resistor. Soft-start setting Figure 4 shows the typical soft-start timing waveforms of RT9206. The soft-start time of Buck converter can be set by selecting the soft-start capacitance value. The delay time between input voltage applied and output voltage starting to ramp up (TDELAY) is calculated as: The total time from input voltage applied to output voltage buildup (TVR) is calculated as :
TVR = 57 x CSS x 10
6
(ms) (13)
When VSD >300mV, the current limit function will be activated and latch the controller. So the current limit function can be set by MOSFETs selection. The relation of maximum inductor current IL(LIM) and on-state resistance of MOSFET (RDS(ON)) is described as : -3 300 x 10 () (11) RDS(ON) = IL(LIM) Setting the Output voltage The output voltage is set by external voltage divider and reference voltage. The feedback pin (FB, LFB1, and LFB2) is connected to the inverting input of error amplifier and is referenced to 0.8V reference voltage at non-inverting input as shown in Figure 3.The output voltage is set by the following equation.
VOUT = (1+ Ra Rb
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The effective soft-start time (TSS) during that output voltage ramps up from zero to set voltage is calculated as :
VOUT VIN
TSS = (320 x
) x 10 x CSS
6
(ms) (14)
Besides, appropriate soft-start capacitor should be selected so that the start-up current will not trigger the current limit function. And make sure that the input power source could supply the soft-start current. The total time from input voltage applied to power good signal pull-high (TPGOOD) is calculated as :
TPG = 640 x CSS x 10
6
(ms) (15)
) x 0.8
(12)
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RT9206
VDD VSS
Boost Component Selection The booststrap gate drive circuit is used to drive high side N-channel MOSFET. The boost capacitor should be a good quality and can operate in high frequency. The value of boost capacitor depends on the total gate charge (QHg) to turn on the MOSFETs. Assuming steady state operation, the following equation can be used to calculate the capacitance value to achieve the targeted ripple voltage VBOOT .
CBOOT = QHg VBOOT
VOUT
TVR
PGOOD
TSS TPGOOD
(F)
Figure 4. The soft-stat timing diagram of RT9206 For the example of CSS = 1F, VIN = 12V, VOUT = 5V, then TVR = 57ms, TSS = 133ms and TPGOOD = 640ms. Shutdown The power stage can be shutdown by pulling soft-start pin below 0.7V. During shutdown, both of high side MOSFET (S1) and low side MOSFET (S2) are turned off. Setting the switching frequency The switching frequency can be set by a resistor (RRT ) connecting between RT and GND pins. Equation (16) describes the relationship of RRT and switching frequency. As RT open the normally operated frequency is 200kHz. 62 x 10
8 3
The capacitor in the range of 0.1uF to 1uF is generally adequate for most applications. The VINT pin bypass capacitor CINT needs to charge the boost capacitor, to drive the low side MOSFET, and to power the RT9206. CINT should locate near VINT and GND pins with short and wide traces. Generally, a 4.7uF high frequency ceramic capacitor is recommended. Feedback Compensation The RT9206 is a voltage mode controller. The control loop is a single voltage feedback loop including a transconductance error amplifier and a PWM comparator. To achieve fast transient response and accurate output regulation, appropriate feedback compensation is necessary. The goal of the compensation network is to provide a closed loop transfer function with the highest 0dB crossing frequency and adequate phase margin. Generally, the phase margin in a range of 45 ~ 60 is desirable. Figure 4 shows the simplified diagram of synchronous buck converter and control loop.
RRT =
fS - 200 x 10
()
(16)
RRT Connecting Between RT and GND Pins fS(kHz) 250 300 350 400 450 500 550 600
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RRT (k) 120 55 37.5 30.6 24.4 22.5 19.3 16.8
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RT9206
iS1 L + VL S1 S2 iS2 + VOR + VOC iL
iC rC RL
+
IOUT +
Next, deriving the transfer function d(s)/vC (s) of the direct duty ratio pulse-width modulator (PWM Generator). The transfer function Tm(s) of the modulator is given by Tm(S) = d(S) 1 = VC(S) Vr (19)
VIN
VOUT
COUT -
where, Vr is the amplitude of ramp-waveform which is listed in datasheet.
Sensor Gain
Compensator d + PWM Generator VC Cc2 Rc1 Cc1 + gm -
VREF
For simplification, the transfer function of PWM generator and Buck converter can is combined. The resulting is shown in equation (20)
G(S) = VOUT(S) = VC(S) VIN 1 + rc x CO x S x L Vr S x L + CO x S( + rc x CO) + 1 RL (20)
2
Ra
Rb
The transfer function of Equation (20) is a second order system and Bode plot is shown in Figure 7.
Gain
Figure 5. The simplified diagram for synchronous Buck converter and control loop. From control system point of view, the block diagram of Figure 5 is shown in Figure 6.
VIN/Vr
fp
G(S) Compensator VREF + C(s) Vc(s) PWM Generator 1/Vr d(s) BUCK Converter Gp(s) VOUT
f fz
Phase
0
Rb Ra+Rb Sensor Gain H (s)
f
-90 -180
Figure 6. The control block diagram of synchronous Buck converter
Figure 7. The Bode plot of Buck power stage
First, deriving the accurate small-signal models of power stage, the equation (18) is the transfer function of vO(s)/d(s), which be obtained by space averaging technique.
VOUT(S) GP(S) = = d(S) 1 + rc x CO x S x VIN L S 2 x L x CO + S( + rc x CO) + 1 RL (18)
In Figure 7, the resonance of the output LC filter produces a double pole and -40dB/decade slop. The resonance frequency is expressed as follows : fP = 1 2 x L x CO (Hz) (21)
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RT9206
The effective series resistance (ESR) of capacitor and capacitance introduces one zero into system, the zero is given as : 1 (Hz) (22) fZ = 2 x rc x CO In the voltage-mode Buck converter shown in Figure 5, the loop gain of system is 1 TL(S) = C(S) x x GP(S) x H(S) = C(S) x G(S) x H(S) (23) Vr The desired loop gain and phase margin is show in the Bode plot of Figure 8. Where the f C is zero crossover frequency defined as the frequency when the loop gain equals unity. Typically, fC be chosen in range 1/10 ~ 1/20 of switching frequency. f C determines how fast the dynamic load response is. The higher f C with the faster dynamic response, and the phase margin in the range of 45 ~ 60 is desirable. So, the transfer function of compensator C(s) must be designed to meet these requirements. In many applications, use an electrolytic capacitor as the output capacitor, if the zero (f Z) caused by effective series resistance (ESR) of capacitor is a few kHz and smaller than 8 times f P, the type 2 (PI) can be used to get desired compensation. Figure 9 shows the typical type 2 trans-conductance error amplifier and the Bode plot is also shown in Figure 10.
VOUT
Power stage VIN/Vr
G(s) Gain
VREF
Ra
+ gm -
Vc
Rc1 Rb
fp f fz
Cc2
Cc1
TL(s) Desired loop gain
Figure 9. The typical type 2 trans-conductance error amplifier.
fc
f Gain(dB)
Phase gmRc1 0 f f Phase fcz fcp
-90 Phase -180 margin Boost -90 f
Figure 8. The Bode plot of desired loop gain and phase margin
Figure 10. The Bode plot of type 2 trans-conductance error amplifier
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DS9206-11 March 2007
RT9206
The design procedure as following : (1). Selecting the zero crossover frequency f C is 1/10 ~ 1/20 switching frequency. Then according equation (24) set the resistor RC1 to determine the zero crossover frequency. Vr x L x f C V OUT () (24) R C1 x V IN x gm x r C V REF (2). Place the zero of compensator is 70% fp that is resonance frequency of power stage. The compensator capacitor Cc1 can be selected to set the zero. The equation is shown in following : L x CO 0.7 x RC1 Step2. Determine the zero crossover frequency and compensated type. Select desired zero-crossover frequency :
fC fS/10 ~ fS/20
Select f C = 20kHz Step3. Determine desired location of poles and zeros for type2 compensator. Select:
fCZ = 0.7 x fP = 0.7 x 1.34kHz = 938Hz
Assume
fCP =
fS 2
= 100kHz
CC1 =
(F)
(25)
Step4. Calculate the real parameters-resistor and capacitors for type2 compensator. From equation (21), the RC1 is calculated as following : f C x L x Vr rC x V IN x gm V OUT V REF x 5V 0.8V = 8.4k
(3). Set a second pole to suppress the switching noise. Assume the pole is one half of switching frequency f s, which results in capacitor Cc2 as shows in following:
CC2 = 1 x RC1 x fs 1 CC1 1 x RC1 x fs
RC1 =
x
(F) (26) =
20kH z x 15 H x 1.9 22m x 12V x 1.6m s
Design example Design example of type 2 compensator: the schematic is shown in Figure 4, where the parameters as following: V IN =12V, V OUT =5V, I OUT =5A, switching frequency=200kHz, L=15H, CO=940F, rC=22m, the parameters of RT9206 as following: gm=1.6ms, ramp amplitude=1.9V,and reference voltage Vref=0.8V. Step1. Determine the power stage poles and zeros. The pole caused by the output inductor and output capacitor is calculated as : fP = 1 2 L x CO 1 2 x rC x CO = 1 2 15 x 940 1 2 x 22m x 940 F = 1.34kHz Select RC1 = 8.2k Calculate CC1 from equation (25) CC1 = L x CO 0.7 x RC1 = 15 x 940 0.7 x 8.2k = 20.7nF
Select CC1 = 22nF Second capacitor CC2 can be calculated using equation (26) CC2 = 1 1 = = 194pF x R C1 x fS x 8.2k x 200kHz
fZ =
=
= 7.7kHz
Select CC2 = 220pF
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RT9206
Linear Regulator
Output Capacitor Selection Solid tantalum capacitors are recommended for use on the output capacitors of LDO because their typical ESR is very close to the ideal value required for loop compensation. Tantalums also have good temperature stability: a good quality tantalum will typically show a capacitance value that varies less than 10-15% across the full temperature range of 125C to -40C. ESR will vary only about 2X going from the high to low temperature limits. Linear Regular MOSFETs Selection The main consideration of pass MOSFETs of linear regulator is package selection for efficient removal of heat. The power dissipation of a linear regulator is Plinear = (VIN - VOUT) x IOUT (W) (26) (1). The IC needs a bypassing ceramic capacitor C1 as a R-C filter to isolate the pulse current from power stage and supply to IC, so the ceramic capacitor C1 should be placed adjacent to the IC. (2). Place the high frequency ceramic decoupling close to the power MOSFETs. (3). The feedback part should be placed as close to IC as possible and keep away from the inductor and all noise sources. (4). The components of bootstraps (C8, C9 and D1) should be closed to each other and close to MOSFETs. (5).The PCB trace from UGATE and LGATE of controller to MOSFETs should be as short as possible and can carry 1A peak current. (6). Place all of the components as close to IC as possible. Figure 11 shows the typical PCB layout of synchronous Buck converter with RT9206 controller
The criterion for selection of package is the junction temperature below the maximum desired temperature with the maximum expected ambient temperature. Layout Consideration Layout is very important in high frequency switching converter design. If designed improperly, the PCB could radiate excessive noise and contribute to the converter instability. First, place the PWM power stage components. Mount all the power components and connections in the top layer with wide copper areas. The MOSFETs of Buck, inductor, and output capacitor should be as close to each other as possible. This can reduce the radiation of EMI due to the high frequency current loop. If the output capacitors are placed in parallel to reduce the ESR of capacitor, equal sharing ripple current should be considered. Place the input capacitor directly to the drain of high-side MOSFET. The MOSFETs of linear regulator should have wide pad to dissipate the heat. In multilayer PCB, use one layer as power ground and have a separate control signal ground as the reference of the all signal. To avoid the signal ground is effect by noise and have best load regulation, it should be connected to the ground terminal of output. Furthermore, follows below guidelines can get better performance of IC:
Figure 11. The PCB layout of synchronous Buck converter with RT9206 controller
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DS9206-11 March 2007
RT9206
Outline Dimension
A
H M
J
B
F
C I D
Symbol A B C D F H I J M
Dimensions In Millimeters Min 9.804 3.810 1.346 0.330 1.194 0.178 0.102 5.791 0.406 Max 10.008 3.988 1.753 0.508 1.346 0.254 0.254 6.198 1.270
Dimensions In Inches Min 0.386 0.150 0.053 0.013 0.047 0.007 0.004 0.228 0.016 Max 0.394 0.157 0.069 0.020 0.053 0.010 0.010 0.244 0.050
16-Lead SOP Plastic Package
Richtek Technology Corporation
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Richtek Technology Corporation
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